EP 1 548 934 B1

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EP

1

548

934

B1


(12)

EUROPEAN PATENT SPECIFICATION


(45)

Reference to the grant of patent:


03.09.2008


Patentblatt

2008/36


(21)

Application number:

04030390.1


(22)

Date of filing:


22.12.2004


(51)

Int. Class:

H03F 5/00


(2006.01)

H03F 1/32


(2006.01)

H03F 3/26


(2006.01)




(54)

Vollsymmetrischer Leistungsverstärker

Fully differential push
-
pull amplifier

Amplificateur push
-
pull totalement différentiel


(84)

Named
Parties:


AT BE BG CH CY CZ DE DK EE ES FI FR GB GR
HU IE IS IT LI LT LU MC NL PL PT RO SE SI SK
TR


(30)

Priority:

22.12.2003

DE

10360347


(43)

Of publication of application:


29.06.2005


Patentblatt

2005/26


(73)

Proprietor:

Blöhbaum, Frank


79112

Freiburg i.Br.

(DE)


(72)

Inventor:




Blöhbaum, Frank

79112 Freiburg i.Br.

(DE)


(56)

References cited: :

WO
-
A
-
00/11779

US
-
A
-

4 229 706

US
-
B1
-

6 242 977

FR
-
A
-

2 547 470

US
-
A
-

4 531 100











Note: Within nine months from the publication of the mention of the grant of the notice to the European Patent Office of oppo
sition to the European patent granted an appeal. The appeal must be
submitted in writing and justified. It shall not be inserted as

if the opposition fee has been paid.
(Art. 99 (1) European Patent Convention).



Description


[0001]


The main quality criteria for power amplifiers, particularly for applications in audio,
are the lowest possible distortion, high bandwidth, the lowest
possible internal resistance
as low transient intermodulation distortion and high stability of the operating point of the
power components.


[0002]


Power amplifier can be constructed with very different active components whose
properties generally determi
ne the structure of the amplifier: tubes of different construction,
forAs triodes, tetrodes, pentodes, or semiconductor devices, suchAs bipolar transistors,
field effect transistors, MOSFETs and IGBTs. Tubes have to be used as power amplifiers
usually have

a relatively high internal resistance and therefore must be adjusted with the
aid of specially constructed transformers, the transformers, the low impedance of the
speaker. These transformers, however, limit the achievable bandwidth and are themselves
maj
or source of nonlinear distortion. Which therefore desirable direct coupling of the low
-
impedance load, as a rule of the loudspeaker requires the use of active components with
the lowest possible internal resistance or as large as possible steepness. For t
his reason,
as the art enforced the use of semiconductor devices such as bipolar transistors,
MOSFETs, IGBTs, or in rarer cases.


[0003]


Particularly frequently be realized amplifier with higher power requirements push
-
pull amplifier as complementary, whe
rein complementary bipolar transistors (npn and pnp
transistor) or complementary MOSFETs (n
-
channel and p
-
channel transistor) or
complementary IGBTs (n
-
channel and p
-
channel type) with a ground potential related to
bipolar or unipolar voltage source are us
ed.


[0004]


The common feature of in Figures 1 and 2 is illustrated the principle of a typical
semiconductor
-
tipped push
-
pull amplifier or a corresponding push
-
pull output stage is the
use of a complementary transistor pair. For closed
-
circuit setting via

the complementary
transistor pair serves an adjustable bias voltage which is between the two base terminals
of the complementary transistor pair. The other circuit components are then different from
each other.


[0005]


In the basic circuit of Figure 1 ar
e used in the power unit at least two series
-
connected voltage sources whose junction point is connected to ground, so that, based on
the ground potential is a positive and a negative supply voltage. The drive signal is related
to ground potential balanced

signal. The load, suchExample of the loudspeakers, is
located between the connection point of the complementary transistor pair and the ground
terminal.


[0006]


In the basic circuit of Figure 2 in the power supply only a single voltage source is
used wit
h one
-
sided ground connection. The drive signal is a unipolar signal related to
ground potential. If the load comprises a DC path, it must capacitively from the lying on
different levels output terminals, which are formed by the junction of the complementa
ry
transistor pair and the ground terminal, are separated.


[0007]


At the latter feature is also no significant changes by the insertion of a very low
resistance measurement between the load resistor and the speaker and the mass, as
suggested in some publ
ications for current measurement.


[0008]


Are complementary push
-
pull amplifiers with tubes, it is not, as control valves
according to the flow of electrons between the cathode and anode, only a negative power,
so that a lack of complementary electron tub
e amplifier element.


[0009]


For the realization of highly linear power amplifier, the use of tubes as a power
amplifier would be very desirable because they can be used in this application because of
its square tube current
-
voltage characteristic curve v
ery low distortion. The ideal would be
the combination of tubes as a voltage amplifier with high share semiconductor devices,
such as bipolar transistors, MOSFETs or IGBTs, as output
-
side power amplifier. Of great
advantage here would be a dc coupling betw
een the gain stages, so that would otherwise
be required to provide isolation for coupling or bypass capacitors or transformers have no
detrimental effect on the frequency response and harmonic distortion.

Developments to date have focused on the optimizat
ion of pure solid state amps on the
one hand and pure tube amplifiers on the other. In addition, are also occasionally known
amplifier with hybrid assembly components, separated by pipes or Transistor optimized
using pre
-
and power amplifier included, which

are usually capacitively coupled to each
other.


[0010]


When the complementary push
-
pull output stages of FIG 1 with the two opposite at
ground potential related polarity voltage sources, the use of complementary active
components in the power unit neces
sarily inherent to a non
-
symmetric structure. Thus,
both branches of the push
-
pull amplifier are not working up to the same devices, and thus
in principle only partially symmetric. The full electronic symmetry would require that up to
the line type all the

electronic properties of npn and pnp transistors are equal. This is only
approximately attainable. The differences disturb the symmetry behavior of the gain in the
two signal paths and others complicate the realization of further distortion reducing circu
it
structures.


[0011]


A problem in search of a possible symmetrical complementary transistor pair is the
very limited selection of pnp bipolar transistors, p
-
channel MOSFETs and p
-
channel IGBTs
for high power amplifiers. This issue is not resolved by the

creation of a quasi
-
complementary output stage, but moved at the expense of symmetry only in the driver
stage. The complementary pairs of transistors used in the real world are not real couples
on closer analysis. Thus, differing complementary transistors

of a complementary pair, for
example, substantially in terms of the amount of current amplification factor in bipolar
transistors or the steepness with MOSFETs and IGBTs. This is a major cause of non
-
linearities in the amplifier behavior.


[0012]


For exa
mple, in the same drain
-
source withstand voltage, the same maximum
allowable drain current and the same maximum permissible power loss of the chip size of
p
-
channel MOSFETs approximately three times larger than the chip area of
the
complementary n
-
channe
l MOSFETs. This inevitably leads to considerably different
capacities, particularly the gate
-
source capacitance and the drain
-
source capacitance. Due
to the different capacitive loads result in problems of control and no further circuit
measures result in
different slopes (= Slew
-
rate) for positive and negative edges at the
amplifier output. If such a negative feedback amplifier, the frequency compensation is
determined by the much larger capacity of the negative branch (p
-
channel MOSFETs and
IGBTs). This m
ay reduce the achievable bandwidth performance unacceptable degree or
cause instability. In bipolar transistors, these Anpassprobleme (= matching problems)
which are necessary for the complementary symmetry, similar.


[0013]


In summary, it should be noted

that inherently to the prior art push
-
pull amplifier in
complementary technology can not achieve full symmetry. Several methods are known
which attempt to rectify the resulting non
-
linearities by means of a negative feedback and /
or a forward error corre
ction (= Forward Error Correction). The required amount of circuitry
for high quality power amplifier is very high. Furthermore, the correct operation of the error
compensation of the typical applications for audio control with transient pulses critical. F
or
example, in the

US Patentanmeldung US 5,892,398
a power amplifier described by the
output signal controlled operating voltages used in accordance with the bootstrap principle
for the compensation module for which highly linear production is again difficult. The
typical audio power amplifier case, the co
ntrol complex, and capacitive loads such as
speakers and crossovers of different designs, requires a high degree of stability in the
negative feedback condition, the conventional complementary push
-
pull output stages due
to the different high
-
frequency pro
perties of the two signal paths is hardly available.


[0014]


Transformerless push
-
pull power amplifier, use the active components only one
conductivity type and the strictly symmetrical with respect to the electrical properties can
be realized have so far

been particularly associated with tubes on the market. They are
commonly referred to as OTL amplifier (Output Transformer
-
Less) and sold commercially
under that name. Tube amplifiers that are built on this principle, have the following
fundamental disadva
ntages: tubes have too little slope to the low impedance of the load
resistor and the speaker directly control. Without further circuit measures will result from a
small deliverable power and a very high total harmonic distortion.


[0015]


To increase the
steepness, many tube systems are connected in parallel. This can
be excessive in parallel tubes or Tube systems, the slope of the resulting power
component to a value just sufficient to increase. The distortion factor is still high. There
are, however sign
ificant problems with the stability of the operating points of the many tube
systems. The thermal drift of the anode current at a given bias and the variation of the tube
characteristics due to aging, such as eg declining Katodenemissionsfähigkeit lead, at

best,
to an increased distortion, in the worst case, the total failure of the device.


[0016]


The operating point of the power stage is from these tube circuits using a fixed grid
bias such as in

US
-
Patent 4,719,431
or set each by a cathode resistor. The
associated pre
-
amplifier and the voltage is capacitively coupled. Examples of this are shown in U.S.
Patents

US 4,719,431
in Fig.4 and

US 6,242,977 B1
in Fig.2. A DC coupling is in
accordance with the prior art only indirectly by means of a high
-
resistance v
oltage divider,
which in turn are bridged capacitively must. Without this bypass capacitor of the AC
voltage drop would be much too large. On the other hand a direct DC coupling the drift
was already highly vulnerable operating point of the output tube or
the parallel
-
connected
output tubes would destabilize further, as would then affect the drift of the Vorröhre
multiplied with the DC gain of the output tube (s) in addition.


[0017]


According to the state of the art in tube circuits inevitable use of
coupling and
impair / or bypass capacitors, the achievable linearity significantly. As

H. Lemme in the
"Electronics" issue 10/2003, pp. 90
-
94with the article "capacitors as troublemakers"
increase, particularly capacitors demonstrate a high DC voltage stre
ss on high quality amp
distortion to the unacceptably high levels.


[0018]


In the

US
-
Patent US 4,229,706
is a dc
-
coupled push
-
pull amplifier in
semiconductor technology, the output stages consist of two identical npn power
transistors, which are fed by two

"floating" voltage sources. The two bipolar transistors of
the output stage are cross
-
coupled with respect to their load output. The pre
-
amplifiers for
each signal path via a respective driver circuit having a differential input. To the two non
-
inverting
inputs the signal to be amplified is applied as a differential voltage. The two
inverting inputs are connected via a resistor network with the two load terminals and with a
negative pole of a mass
-
based bias. The outputs of the two driver circuits each fee
ding
directly the base of the associated output transistor.


[0019]


In

US 4,229,706
Although widely described arrangement is symmetrical and DC
-
coupled, but this is paid for by the following disadvantages:


[0020]


The kind of realized quiescent current se
tting works only with bipolar transistors,
since the height of the base current (current flow of the base
-
emitter junction) on the driver
and the resistor network is controlled.


[0021]


The quiescent current setting of the exclusive control of the base cu
rrent works in
cross
-
coupled amplifiers only using the same ideal of bipolar power transistors. In reality,
the inevitable spread of the current amplification factor absolute and, in dependence on
the temperature in significant differences between the quie
scent currents of the two power
transistors. For an unacceptable high false differential current caused by the load
resistance of the speaker.


[0022]


Moreover, the exclusive use of semiconductor components in the circuit of the

US
-
Patent 4,229,706
therefo
re also a disadvantage, since then the desirable flexibility of the
design is very limited. Thus, for example, extremely difficult to realize with a low
-
distortion
power amplifier tubes, which can drive the high base currents of the bipolar power
transisto
rs which can be used exclusively. It would be desirable in this case the use of
MOSFETs or IGBTs as power device, because these devices are voltage
-
controlled and
therefore a combination with low distortion power amplifiers on tube base would be
possible.


[0023]


In

US 6,242,977 B1
is in Fig.4 a push
-
pull amplifier is shown with only one
conductivity type MOSFETs voltage amplifier without a preceding, wherein the two
MOSFETs via two "pseudo" floating voltage sources are fed. The two spurious floating
voltag
e sources are formed here by means of a capacitive decoupling from a single
voltage source. This circuit type has practically realizable high quality power amplifier
following disadvantages:


[0024]


The operating point of the MOSFETs by means of a fixed
gate voltage via a
voltage divider on the one hand (cf. FIG 4, node 276, resistors 308 and 310) and a control
voltage (from the Regleinrichtung 268) on the other set. The temperature
-
dependent
change of the drain current at constant gate
-
source voltage res
ults in the worst case, a
significant change in the absolute value of the drain current, and can even lead to thermal
destruction of the MOSFET. The set DC gate voltage must always be the outcome of the
potential upstream but in none of the

US 6,242,977 B1
embodiments shown by way of
capacitor are electrically isolated. This is the desirable realization of a DC
-
coupled
amplifier without Koppe Lund / or bypass capacitors is not possible.


[0025]


In 4 of

US 6,242,977 B1
shown generation of floating voltages fr
om a single voltage
source for power amplifiers problematic because there occur, may require highly volatile,
high current small decoupling resistors 286, 288, 290 and 292 and then in succession
extremely large capacitors 294 and 296. These capacitors omit
ted if the two power
amplifiers in each case a real floating voltage source as in the mentioned prior art in FIG 1
of
US 6,242,977 B1
is provided.


[0026]


The object of the invention, the indication of an improved push
-
pull amplifier having
a fully symmetri
cal structure with functional units of the same conductivity type, is as large
as possible freedom for the use of tubes and / or semiconductors and / or monolithic
integrated circuits permits a transformerless connection enables the load and on coupling
wa
ived and bypass capacitors and transformers.


[0027]


The problem is solved by the features of claim 1 The advantages of the invention
are that the improvements proposed a power amplifier is realized with very high linearity,
preferably as audio amplifier
is of the highest quality category. The operating points are
adjustable in a simple manner, wherein residual asymmetries of the functional units via
additional control devices are securely stabilized. The control devices can be formed so
that they can comp
ensate for changes due to drift and temperature. By the required
flexibility can be used for the power amplifiers and amplifier blocks in each case the most
appropriate components and combine them. Whether the power amplifier is basically
driven by a balan
ced differential signal to the ground reference level or whether the
difference signal is a DC level is superimposed on, or whether a unipolar drive signal is
present, at most minor changes required in the two signal paths or in the control
equipment.


[00
28]


The invention and advantageous embodiments and developments are now
described with reference to the accompanying drawings:

Figures 1 and 2 schematically show two well
-
known complementary amplifier,

3 shows schematically a known push
-
pull amplifier wit
h identical and DC
-
coupled
functional units and with cross coupling in the power stage,

4 shows schematically a first embodiment of the invention,

5 shows schematically an embodiment of an offset voltage control,

6 shows an embodiment with an extended offs
et voltage control,

7 shows an embodiment with tubes in the power stage,

8 shows an embodiment with a different bias current setting,

9 shows an embodiment with modified input terminals,

10 illustrates an embodiment with optocouplers for quiescent current
setting and

11 shows an embodiment with an asymmetrical signal input.



[0029]


The well
-
known complementary amplifier of Figure 1 contains in series circuit
comprises an NPN transistor Q1 and a pnp transistor Q2 between two voltage terminals +
Ub
-
Ub and b
y two series
-
connected voltage sources V2, V3 are supplied. The connection
node of these voltage sources is connected to the ground potential GND and to a terminal
of the load RL. The other terminal of the load RL is located at the connection node of the
t
wo transistors Q1, Q2. For the quiescent current setting of the two transistors Q1, Q2
provides an adjustable bias voltage V bias between the two bases of these transistors. By
suitably adjusting the bias voltage Vbias is the bias current can be set betwee
n the two
transistors so that they, forIn each instance, AB amplification operation work. The single
ended input voltage is supplied via an input + in the base of the npn transistor. The
amplifier is complementary with respect to the mass away from load te
rminal non
-
inverting. Capacitors for electrical isolation are not shown in Figure 1. They are in any case
necessary if the input signal is a DC voltage is superimposed and the
Eingangsspannungshub is not very large Q1 relative to the base
-
emitter voltage o
f
transistor.


[0030]


The well
-
known complementary amplifier of Figure 2 requires in contrast to FIG 1,
only one voltage source V4 for the two Q3 in series connected complementary transistors,
Q4 from an NPN and a PNP transistor, which lie between a posit
ive supply terminal + Ub
and the ground terminal GND . The voltage source V4 is connected at one terminal
connected to ground GND and the other fed with the supply terminal + Ub. The quiescent
current adjustment for the two transistors is carried out as in

Figure 1 with an adjustable
bias voltage Vbias, which lies between the two base terminals. Since the common emitter
terminal is attached to this complementary circuit also approximately not more at the
ground potential GND, there is in the idle state, a v
oltage difference between the common
emitter terminal and the ground GND, which corresponds approximately to half the supply
voltage V4. Be the to be connected between these two nodes must load RL, if it has a
direct current path, therefore separated from
this DC voltage through a coupling capacitor
C2. In addition to this coupling capacitor C2, the signal input + are also through a capacitor
separated in voltage from the signal source, since the signal input + has approximately
likewise half the supply vol
tage V4 and thus generally very different from the rest potential
of the signal source is different.


[0031]


3 shows the already mentioned in

US
-
Patent US 4,229,706
Audio
-
described push
-
pull amplifier having a symmetrical structure, which is in the two
signal paths DC
-
coupled
and whose two npn output stage transistors Q1, Q2 respectively from a floating voltage
supply V1, V2 are fed. The isolation of the ground reference potential GND is achieved
with the floating power supply usually uses its own power
transformer and a bridge rectifier
circuit. The load RL is connected between the two emitters, which also form the bases for
the connected crosswise floating supply voltages V1, V2. The balanced input signal is
supplied via the inputs in + and
-
in, wherein
the positive input + in non
-
inverting the input of
a is executed in bipolar operational amplifier U1. The negative input
-
in is connected to the
noninverting input of a first operational amplifier U1 identical second operational amplifier
U2. The output of
the first and second operational amplifier is directly, ie without coupling
capacitor, connected to the base of the first and second output transistors Q1, Q2
connected. The power supply of the two V3 is operational amplifier via two series
-
connected volta
ge sources, V4, whose point of attachment to the ground potential GND is
connected and operational amplifier whose mass facing away from poles, the positive and
negative supply voltage + V,
-
V for both U1, form U2.


[0032]


The quiescent current adjustment of the two output transistors Q1, Q2 via a
respective resistor R4 and R5 between the emitter and the negative supply voltage
-
V. The
associated base current, the output of the associated operational amplifier U1 and U2.
He
re now is a problem, because by the resistance R4 and R5 certain current must be
identical to the operational amplifier U1 and U2 be supplied base current, since this current
V3 only with the voltage sources, V4 is linked and no other current path is avail
able. It is
therefore not regulated by the emitter bias current, but the base current. The resulting
emitter bias current, the substantially from the floating voltage supply V1 or V2 is fed and
flows back through the load RL is, directly from the current a
mplification factor of the
respective output transistor Q1 and Q2
-
dependent and sprinkled with what is actually
undesirable. The current regulation, strictly speaking, the control of the base current is
performed by tapping the respective emitter potential

and return on a relatively high value
resistor R2 and R3 to the inverting input of the first and second operational amplifier U1,
U2. Thus both emitters that give rise to the termination points of the load RL, as have the
same potential, these two circuit

nodes also has a relatively high value resistor R6 is
connected and thus is the difference between the two potentials as a kind of offset voltage
at the inverting inputs of the operational amplifier U1, U2 returned. Thus in the control
case, the relativel
y small quiescent current does not hinder the modulation of the output
transistors Q1, Q2, the respective emitter current increased by a resistor
-
diode path R7,
D1 and R8, D2, which represents the respective emitter niederohmigeren a current path to
ground

GND available when the diode is conducting. It is mentioned that by the shown in
Figure 3 base current return to the operational amplifier U1, U2 only current
-
controlled
amplifiers, bipolar transistors therefore be able to be controlled, but not voltage
-
c
ontrolled
components.


[0033]


4 shows schematically the circuit diagram of a first embodiment of a symmetrical
push
-
pull amplifier according to the invention. The two signal paths are symmetrical and
each contain a Vorverstärkerblock N1B or N2B and an out
put amplifier and M1 M2. The
power amplifier M1, M2 are connected to a floating voltage supply V1, V2, wherein the
load circuits through a load RL, which between the base points of the two amplifiers is M1,
M2 are connected crosswise closed. The two base p
oints of the amplifier is set at the
same time, the connection points or terminals RL1, RL2 for the externally connected load
RL and form the output signal.


[0034]


The power supply device of the amplifier blocks N1B, N2B contains at least two
series
-
conn
ected voltage sources V3, V4, whose junction point is connected to ground
GND. The resulting supply voltages + Ub
-
Ub and feed the supply terminals of the amplifier
blocks that exist in the embodiment of FIG 4 each from an input
-
side amplifier N1, N2 with
h
igh open
-
loop gain. The positive pole of the output stage of the amplifier N1, N2 is fed to
the positive supply voltage + Ub or with a different positive and also referenced to ground
voltage source. The negative pole of the output stage of the amplifier N
1, N2 is connected
via a potential difference producer 4, 5 RL1 associated with the load connection, RL2
connected. 5 About this potential difference generator 4, an impressed current flows. Due
to the high open loop gain in conjunction with a high impedan
ce non
-
inverting and
inverting input high impedance, with negligible offset voltage and negligible offset current
and with a low impedance output, the amplifier N1, N2 conduct such as a more or less the
ideal operational amplifier.


[0035]


As a power ampl
ifier M1, M2 are shown in FIG 4, two three
-
pole, whose respective
input electrode to the output of the preceding amplifier N1, N2 is connected. The first is M1
power amplifier is of a floating first voltage source V1 and the second amplifier M2 fed
from a
floating second voltage source V2. The first is M1 amplifier via its first supply
terminal connected to a pole V1 of the first voltage source and via its second supply
terminal, which also serves as Endverstärkerausgang, RL1 to the first output and the loa
d
RL to the other pole of the first voltage source V1. The second amplifier M2 is via its first
supply terminal with a pole of the second voltage source V2 and via its second supply
terminal, which also serves as Endverstärkerausgang, to the second output
RL2 and via
the load RL to the other pole of the second voltage source V2,


[0036]


The required operating
-
point determining control voltage is in the corresponding
potential difference generators 4, 5 by a determined by a Ruhestromeinstelleinrichtung Ix
w
ith an adjustable current source I1 predetermined current that generates the potential
difference producer 4, 5 the desired potential offset. The negative to the supply terminal
-
Ub connected Ruhestromeinstellei nrichtung Ix contains an adjustable current so
urce I1,
the current means of the two to the output terminals RL1, RL2 connected resistors R2, R3
is split equally between the two potential difference generator 4, 5, so that the operating
point for the two amplifier M1, M2 is symmetric.


[0037]


This adj
ustment can of course be done in different ways, for example, purely
manually as in the embodiment of FIG 4 on an adjustable current source I1, an adjustable
voltage source V bias (see Figure 8) or via a potentiometer, which controls the current
source I1.

Expedient is also a temperature
-
influenced control, via a temperature sensor
that detects the temperature of at least one power component, the adjustable voltage or
current source I1 and Vbias so affected that the quiescent current of the amplifier power
components M1, M2 remains constant or has a predetermined temperature profile. Even
more innovative a full automatic control is such that the closed
-
circuit current value of the
power components of the power amplifier M1, M2 is measured and compared with a

predetermined reference value, and any deviations from the desired value by automatically
adjusting the Ruhestromeinstellei nrichtung be controlled to Ix. Of course, combinations of
these control measures are described under protection.


[0038]


The two am
plifier N1 and N2 have an inverting and a noninverting input and owing
to their high open loop gain of a respective operational amplifier. When the resultant output
operational amplifier, unseen by the most appropriate load port RL1 or RL2 of which is
regu
lated by a feedback network to the ground potential GND. The feedback network is
determined by the series connected resistors R6 and R7 or R8 and R9 are formed, which
lie between the load terminal RL1 and ground GND or the load connection RL2 and
ground GN
D. The common junction of resistors R6, R7 and R8, R9 is connected to the
inverting input of the amplifier N1 or N2. N1 is the non
-
inverting input of the first amplifier
to the positive input signal and + in the non
-
inverting input of the second amplifier
N2 is
connected to the negative signal input
-
connected in the push
-
pull amplifier. By the
feedback network R6, R7 and R8, R9, the output of the amplifier N1 or N2 regulated so
that the potential difference generator 4, 5 and the voltage divider R6, R7 and
R8, R9, the
potential at the noninverting input of the amplifier N1 or N2 is identical to the potential at
the positive and negative input signal to + or
-
in. This provision of the feedback network is
independent of the size of the bias current setting for
the performance of the amplifier
components M1, M2, which is determined by the variable current source I1. The resulting
level offset at the potential difference producer 4 or 5 is by the amplifier N1 or N2
corrected.


[0039]


The resistors R10 and R11 fro
m the positive or negative signal input to ground
GND have no significance for the feedback. They serve only to determine the GND
-
potential for the respective non
-
inverting inputs of N1 and N2, and the line matching, since
the signal inputs are high impeda
nce to the noninverting inputs of the amplifier N1 and N2
in the rule. Later, an embodiment is shown in which the signal inputs are connected to the
inverting inputs of the amplifier N1 and N2. With the device connected to the feedback
network inputs, thes
e inputs are then much lower impedance, it can account for the line
adjustment if necessary.


[0040]


In order to realize a fully symmetrical amplifier according to the invention for a
balanced input signal, the corresponding resistors in the two signal br
anches of course, be
equal. It is R6 = R7 = R8 and R9. The ratio of resistors R6/R7 and R8/R9 determines the
overall gain of the balanced amplifier.


[0041]


The upper half amplifier, essentially of the assemblies consisting N1B and M1,
increases the half
-
wave of the symmetrical input signal and the lower half amplifier,
consisting of the modules N2B and M2, reinforces the other half
-
wave.


[0042]


The amplifier M1, M2 can be practically implemented in various ways: as individual
power devices suchAs power
MOSFETs, IGBTs or bipolar power transistors, but also
advantageous from combinations of like or different amplifier components with or without
local negative feedback.


[0043]


In the diagram of Figure 5 is the output stage of the amplifier blocks and N2B
N1B
example as a triode or XTR1 XTR2 executed. In the embodiment shown in FIG 5, the
amplifier blocks N1B and N2B so that each of the combination of an input amplifier N1, N2
with high open
-
loop gain and the amplifier N1, N2 is connected tube XTR1, XTR2, t
he
control grid to the corresponding amplifier output is connected. The anode of the tube
XTR1, XTR2 is connected to the positive supply voltage + Ub or with a different positive
and also referenced to ground voltage source. The cathode of the tube XTR1, X
TR2 is
connected via a potential difference producer 4, 5, which contains the simplest case, a
resistor R4, R5, and over which a load
-
independent current flows, with the associated load
connection RL1, RL2 connected.


[0044]


For the embodiment 5 is shown
as a triode tubes XTR1, XTR2 can of course also
tetrodes, pentodes, and Hexodes Heptoden with the prior art circuit corresponding to the
additional gratings are used.


[0045]


As a power amplifier M1, M2 are shown in FIG 5, two n
-
channel MOSFETs having
the
ir respective gate electrode connected to the cathode of the preceding tube XTR1,
XTR2 is connected. The respective source electrode is connected to the associated load
connection RL1, RL2 and the respective drain electrode to the positive potential of the

floating voltage source V1, V2. The operating
-
point determining gate
-
source control
voltage is in the corresponding potential difference generators 4, 5 by a determined by a
Ruhestromeinstelleinrichtung Ix with an adjustable current source I1 predetermine
d
current, which generates via the resistors R4, R5 the desired potential offset. The negative
to the supply terminal
-
Ub connected Ruhestromeinstelleinrichtung Ix contains an
adjustable current source I1, the current means of the two to the output terminal
s RL1,
RL2 connected resistors R2, R3, R4 equally between the two resistors, R5 divided so that
the operating point for the two MOSFETs M1, M2 is symmetric.


[0046]


To simplify the functional description was according to the embodiment of FIG 4 is
initial
ly assumed that the amplifier N1 and N2 have a behavior like an ideal operational
amplifier. The embodiments from Figure 5 assume that the active components differ in the
two signal branches from the ideal properties and also with respect to the symmetry
c
ondition more or less to each other are different. By means of additional control loops
according to embodiments of the invention can eliminate such shortcomings but for the
overall operation of the push
-
pull amplifier.


[0047]


In the diagram of Figure 5
shows schematically a control arrangement such as the
"bias control" B1 shown that the added push
-
pull amplifier of Figure 4. Since the same
circuit and functional units in all figures of the drawing provided with the same reference
numerals are unnecessar
y, repetitive functional descriptions. The circuit block B1 contains
two control circuits B2, B3, whose inputs are connected to the amplifier outputs RL1, RL2
and their outputs to the feedback networks of the two amplifier blocks N1B or N2B are
linked.


[0
048]


The input of the first control loop B2 is represented by two series
-
connected
identical resistors R16, formed R17, to the outputs RL1, RL2 are connected and their
common connection point avg via a resistor R15 to the inverting input of operational
am
plifier U3 is connected. On the non
-
inverting input of U3, a DC voltage V5 is applied.
The output of amplifier U3 is by means of a capacitor C2 due to the non
-
inverting input.
The power of the amplifier U3 is conveniently carried out via the existing suppl
y potential +
and Ub
-
Ub or another power supply. A at the output of U3 connected inverter N3 inverts
the output signal and supplies it via a resistor R18 back to the common connection point of
the resistors R8, R9 and thus inverting input of the amplifier
N2. For further consideration,
this circuit node, which is an important interface for control and in Fig 5 is shown as a line
labeled "bias 1". A second feedback of the output signal of amplifier U3 is N1 via a resistor
R19 to the common junction of resist
ors R6, R7 and thus inverting input of the amplifier.
For further consideration, this circuit node, which is also represented an important
interface for control and in Figure 5 as a line labeled "Bias 2".


[0049]


The input of the second control loop is fo
rmed by a subtracter B3 D1, whose
minuend input + to the load terminal RL2 and whose subtrahend
-

is connected to the load
terminal RL1. The output of the subtractor D1 determines the potential of the node diff, and
is connected via a resistor R14 to the i
nverting input of an amplifier U4 is connected, which
is connected via a capacitor C1 connected to its amplifier output. This output is also
connected via a series circuit comprising an inverter N4 and a resistor R12 connected to
the circuit node bias 1 an
d via a resistor R13 connected to the other circuit node bias the
second U4 at the noninverting input of the amplifier is connected to the ground potential
GND.


[0050]


The inventive function of the control block bias scheme B1 is that avg means of the
fi
rst control loop B2 the mean of the potentials detected at the outputs RL1, RL2 and
compared with a predetermined reference voltage V5. About the integrator formed by
amplifier U3, the deviation of the potential is increased from the nominal value avg V5.
Means acts of inverter N3 and the resistors R18 and R19 this increased deviation from the
nominal value of V5 in the same way, so the two directly coupled control interface bias 1
and bias 2 that the average of the potentials at the output ports RL1, RL2 t
o the
predetermined target value V5 is regulated. Thus, the amount of the absolute potential of
the outputs RL1, RL2 defined set.


[0051]


Means of the second control loop is B3 diff the time average of the signed
difference detected the potentials at the
outputs RL1, RL2, this value is compared with a
predetermined target voltage and adjusted deviations by means of the amplifier U4 formed
integrator through the two control interface BIAS 1 and BIAS 2 . Since the difference
voltage may be negligibly small a
s a rule is to say in the range of 0 V, is the reference
input (= non
-
inverting input) supplied to the amplifier U4 as a reference voltage, the ground
potential GND. Because possible deviations are usually different direction on the two
control interfaces
bias 1, bias effect 2 must be inserted into the return port on the bias of
an inverter N4.


[0052]


The time constant τ2 = R14 * C1 of the second control loop is B3 selected so that
no interference of signals to be amplified alternating voltage (= AC signa
ls) takes place in
the intended useful frequency range of the balanced amplifier. For use as an audio
amplifier, for example, greater than or equal τ2 1s useful.


[0053]


The time constant τ1 = R15 * C2 of the first control loop B2 is set so that also a
va
lue is greater than or equal achieved 1s in order to avoid the event of any deviations
from the ideal symmetrical behavior to influence the to be amplified AC signals in the
intended useful frequency range.


[0054]


If the amplifier N1 and N2 in the Vorver
stärkerblöcken implemented so that in the
idle state may be considerable, even drifting loaded, the potential difference between the
non
-
inverting and the inverting input occurs this amplifier N1, N2, then by a suitable design
of the bias control of the di
sruptive influence to minimize the operating point stability.


[0055]


Such a case arises, for example, if the amplifier N1, N2 are not executed in
semiconductor technology, but in tube technology. It could, for example, the first stage of a
tube amplifier

can be realized as a cathode base level. In this case corresponds, for
example, the grating the non
-
inverting input of the amplifier N1 or N2 and the cathode to
the inverting input of N1 or N2. By the cathode current results in a rise in potential across
the resistor R7 and R9. Would like in the example of Figure 5, the reference voltage V5 =
0V amount, it would by then idle at 0V potential is governed by absolute and RL1 RL2 a
potential difference across the feedback resistors R6 and R8 arise. This in tur
n leads to a
proportional bias current distribution as a function of the resistors R2, R4, R6 and R3, R5,
R8 with:




[0056]


The operating points of tubes are subjected to thermal and aging
-
related changes.
Thus changes over the temperature and the aging

of the input tubes, which have
substantially the voltage gain in the amplifiers N1 or N2 cause the current component I
(R6) or I (R8), and thus after the first relationship M1 by the I (R4) certain potential of the
gate
-
source path of the output amplifier
. Accordingly, a negative does this by the current I
(R5) certain potential on the gate
-
source path of the amplifier M2. To eliminate this
disturbing influence, or at least minimize, the reference voltage V5 is not fixed, but as the
embodiment of FIG 6, th
e control interface bias 1 and bias 2 on the series
-
connected
equal resistors R24 and R25 linked, so that at the common resistance tap a new node
voltage avgln for a third control loop B4 is available. This node voltage is avgln via a
resistor out R26 to t
he noninverting input of the control amplifier U3, which is different than
not in Figure 5 fed with a reference voltage V5, but of its reference voltage at this terminal
itself generates, connected by a capacitor C3 is there. The average of the potentials
at the
control interface BIAS 1 and BIAS 2 is thus about the RC
-
element low
-
pass filtered R26
and C3 and then serves as a reference voltage U3 at the noninverting input of the control
amplifier. The corner frequency of the RC element R26, C3 is preferably
at least one
decade lower than the lowest selected to be transmitted useful signal. Through this circuit
shown in Figure 6 is the basic version or drift due to the potential difference between the
inputs of the amplifier N1 and N2, on which is recycled via

the negative feedback network
a portion of the output voltage and the output node RL1 or RL2 regulated ideally to zero, or
at least small enough so that the error component of the quiescent current control has no
effect.


[0057]


As already mentioned, an
essential advantage of the proposed by the invention
push
-
pull amplifier principle consists in the largely free choice of the active components.
For example, as power devices, all known types of devices used. In the embodiments of
Figures 5 and 6 were M1 a
s a power amp, M2 showed exemplary n
-
channel MOSFETs.
Without structural changes in these circuits can also use other power devices are used
including those which are not self
-
conducting suchAs IGBTs and bipolar transistors. Be
self
-
conducting components t
hat require a negative bias, for example, used tubes and
some JFETs, can be secured in the same way by a slight modification of the circuit
structure of the invention works. One belonging to this block diagram shows a
corresponding embodiment of FIG 7th


[
0058]


7 shows an illustrative embodiment of the push
-
pull amplifier, power amplifier as
the M1, M2 Xtr3 each tube, which Xtr4 that directly drive the load RL. The amplifier blocks
N1B, N2B included in contrast to the preceding embodiments according to Fig
ures 5 and 6
is not a combination amplifier, but consist solely of the amplifiers N1, N2. These are
implemented so that they Xtr3 at rest, a negative DC output voltage of several volts to
several tens of volts depending on the connected tubes Xtr4 have. Th
e adjustable current
source I1 is connected to this, a positive potential + Ub, so that its owned half current flows
through the resistors R2, R3, the level shifter 4, 5 and the internal output stage of N1, N2
to the negative supply voltage
-
Ub. This curren
t flow caused across the resistors R4, R5
have the required negative grid bias for the tubes or Xtr3 Xtr4. It is thus again Xtr3 with
only one control variable, the quiescent current of both tubes, Xtr4 in the floating solid
mass without reference powered
amplifiers M1, M2 and defines synchronous and
symmetrical set. As previously described, the adjustment of 11 and thus Xtr3 the quiescent
current of the tubes, Xtr4 manually in many ways, are temperature
-
controlled or carried out
by means of an automatic bi
as current control loop. In the exemplary illustration of Figure
7 triodes were used. For Xtr3, Xtr4 can all known types of tubes suchTetrodes and
pentodes as well be used.


[0059]


Does the load RL in a special needs case by means of a transformer on the
equipped with tube output stage M1, M2 are coupled, this can be the same structure as in
FIG done 7, except that then instead of the direct load coupling a transmitter in a known
manner the internal resistance of the tube amplifier to the low
-
load stepped
down. The
transformer can be run cost
-
saving as an autotransformer, because due to the invention
quiescent current setting has no DC bias. Appropriately, the autotransformer winding
symmetrical, for example, in a two
-
chamber assembly, so that the winding r
esistances are
equal for both halves of the winding. Due to the lack of DC bias is a possible realization of
a toroidal transformer. The winding is divided in this case to integer equal parts, which are
then wound in pairs bifilar. A deviation from the sym
metry principle, for example, by simply
winding succession of the respective windings, still allows a stable operation of the
inventive push
-
pull amplifier, exhausted its possibilities, but not fully.


[0060]


A further embodiment of the inventive push
-
pul
l amplifier is shown in FIG 8th This
circuit is almost identical to the circuit of Figure 6, except that the manipulated variable for
the quiescent current adjustment is not carried out as in Figure 6 by the adjustable current
source I1, but by an adjustab
le voltage source Vbias, which ultimately through the
resistors R2, R3 but also shows a generates electricity by means of these resistors R2, R3
is split equally between and through the resistors R4, R5 produced the desired operating
point determined poten
tial offset for the respective gate
-
source junction of the two n
-
channel MOSFETs in the output amplifier M1, M2.


[0061]


According to the invention, the absolute value of the potentials at the load
terminals RL1, RL2 defined and adjusted to a value approx
imately equal. Since the
resistors R2 and R3 are equal, the currents flowing through them are I (R2) and I (R3)
equal to each other. These currents flow almost entirely through the resistors R4 and R5
and the cathode to the anode of the triode or XTR1 XTR2

to the positive supply voltage +
Ub. By the resulting potential offset to the resistors R4 and R5 is a defined positive
potential difference between the gate
-
source path of the MOSFETs M1 and M2, and it
flows logically, a defined quiescent current through

the output stage transistors. The
setting operation of the voltage source Vbias can again take place in many ways, manually
or automatically by means of a temperature
-
controlled quiescent current control loop.


[0062]


According to the embodiment of FIG 9

is an inventive push
-
pull amplifier by the
fact realized that the input
-
side amplifier N1, N2 in the Vorverstärkerblöcken N1B, N2B are
each connected to its inverting amplifier input to the signal inputs
-
in connected or + in the
push
-
pull amplifier. In th
e entrance area of
the amplifier N1, N2 are made only minor circuit
changes. The resistors R7 and R9 is the feedback networks are now not connected to the
ground potential such as in the comparable FIG 5, but to the negative and positive signal
input
-
in
or + in the push
-
pull amplifier. The resistors R10 and R11 to the noninverting
inputs of the amplifier N1 or N2 are at the ends facing away from the amplifier to the
ground potential GND. Further, the recycled lines usually interface bias 1 and bias 2 is n
o
longer as in Fig.5 to the inverting input of the amplifier N1 or N2 recycled, but is non
-
inverting at their inputs. The recycle of the control loops B2 and B3 normally flows through
the resistors R10 and generate R11 the necessary control voltages for th
e amplifier N1,
N2.


[0063]


The embodiment of Figure 10 shows another variation of the bias current
adjustment. The basic circuit of this push
-
pull amplifier is roughly equivalent to the circuit
of FIG 5th The potential difference generator 4 and 5 contai
n, in addition to the resistors
R4 and R5 are each an optoelectronic component or OK1 OK2 or are coupled with such
devices. I1 via a common current source, the light emission is controlled respectively of an
LED element, of the intensity of which the volum
e resistance of a light
-
sensitive photo
-
transistor dependent. This photo
-
transistor is parallel to the already known resistance R4
of Figure 5 or R5, which is also known across the resistor R2 and R3 is energized, the low
end but in contrast to FIG 5 on th
e negative potential Ub is hard. The control of the
potential offset in the resistance R4 and R5 and thus the quiescent current in the amplifier
or M1 M2 (here in each case an n
-
channel MOSFET) now follows by the parallel
connection of the controlled photo
-
transistor distance over which a more or less large
proportion of the resistance R2 and R3 impressed current flows.


[0064]


In the embodiment of FIG 10, the potential difference producer 4 and 5 are thus
formed by an assembly that the resistors R4 and R5

each with an opto
-
coupler or OK1
Trecke OK2 combined in himself. A synchronous setting the quiescent current of both
branches of the floating voltage amplifier is supplied by the application of both optical
coupler with the same control or regulatory powe
r. The power of equality is the
embodiment of FIG 10 in force the current looped the adjustable current source I1 from the
negative supply terminal Ub by both optical coupler OK1, OK2 to the positive supply
terminal + Ub is.


[0065]


The embodiment shown i
n Figure 11 finally shows an advantageous arrangement
of the push
-
pull amplifier according to the invention, where, in principle rather than
balanced input signals only ended input signals to be processed. For comparison, the
circuit is shown in FIG 5th Th
us, both branches of the same signal, ie, at each half cycle to
work may be, the non
-
inverting input amplifier N1 and N2 of the amplifier is controlled at
the inverting input. For this purpose, the non
-
inverting input of the amplifier to the signal
input N
1 + into and through resistor R10 to ground potential GND. The inverting input of
the amplifier is N1 in Figure 4 at the voltage tap of the voltage divider R6, R7, but not the
distal end of resistor R7 in Figure 4 is connected to ground, but to the inverti
ng input of the
amplifier N2 is connected in the second signal branch is. The inverting input of the
amplifier N2 is missing from the 4 known resistor R9, R8 together with the resistance of
the local voltage divider for the feedback from the output RL2. 11

shows this function
assumes the aforementioned resistor R7 by both inverting inputs of both amplifiers N1 and
N2 are coupled to each other. The non
-
inverting input of the amplifier N2 is in contrast to
FIG 4, finally connected through resistor R11 to grou
nd GND. Thus in both branches of the
signal gain is equal to one another, the resistors must follow the following relationship:



[0066]


This provision equation results from the fact that the upper branch as a whole and
non
-
inverting amplifier and the lo
wer branch operates as an inverting amplifier. The above
resistances are therefore not strictly identical, especially for relatively small gains. For
larger gains, the difference in the resistance ratio of pure R6 / R7 or R8 / R7 low.






Claims

1.

Amplifier realised as a symmetrical push
-
pull version having a first amplifying device
and a similar second amplifying device whereat the first amplifying device (N1B, M1) is
connected in a first signal path between a first input (+in) and a first output (
RL1) and a
second amplifying device (N2B, M2) is connected in a second signal path between a
second input (
-
in) and a second output (RL2), furthermore a signal which should be
amplified could be connected to the first input and/or to the second input (+in,

-
in) and an
external connectable load (RL) could be connected to the first and second output (RL1,
RL2),

-

the first input (+in) is coupled to the signal input of a first preamplifying block
(N1B), which is feeded by a voltage supply device (V3, V4), and
whose output is
coupled to the signal input of a first final amplifier (M1) and the second input (
-
in) is
coupled to the signal input of a second preamplifying block (N2B), which is feeded
by a voltage supply device (V3, V4), and whose output is coupled to

the signal input
of a second final amplifier (M2);

-

the first final amplifier (M1) is supplied by a first floating voltage supply (V1) and
the second final amplifier (M2) is supplied by a second floating voltage supply (V2);

-

the first final amplifier (
M1) is connected via its first supply terminal to one terminal
of the first voltage supply (V1) and via its second supply terminal, which is used as
an output of the final amplifier too, to the first output (RL1) und via the load (RL) to
the other terminal

of the first voltage supply (V1), and

-

the second final amplifier (M2) is connected via its first supply terminal to one
terminal of the second voltage supply (V2) and via its second supply terminal, which
is used as an output of the final amplifier too,

to the second output (RL2) und via the
load (RL) to the other terminal of the second voltage supply (V2),

characterized in that

-

a first generator of potential difference (4) between the output of the first
preamplifier block (N1 B) and the first output
(RL1) produces, dependent on the
quiescent current control device (Ix), a first difference voltage between the output of
the first preamplifier block (N 1 B) and the first output (RL1) which determines the
working point and the quiescent current of the fir
st final amplifier (M1) and

-

a second generator of potential difference (5) between the output of the second
preamplifier block (N2B) and the second output (RL2) produces, dependent on the
quiescent current control device (Ix), a second difference voltage

between the
output of the second preamplifier block (N2B) and the second output (RL2) which
sets the working point and the quiescent current of the second final amplifier (M2).




2.

Amplifier in accordance with claim 1,

characterized in that

the first an
d second
preamplifier block (N1 B, N2B) comprise electron tubes and/or semiconductors and on the
output side the voltage alignment to the first and second final amplifier (M1, M2) is done by
means of the first and second generator of potential differences
(4, 5) respectively in
conjunction with the quiescent current control device (Ix).



3.

Amplifier in accordance with claim 2,

characterized in that

additional to the first and
second preamplifier block (N1 B, N2B) a regulating device (B1) dependent on the
offset
voltages at the first and second output (RL1, RL2) applies correction signals (bias2, bias1)
to the first and/or second preamplifier block (N1B, N2B) and therewith said regulating
device (B1) eliminates or at least reduces offset voltages at the fir
st and second output
(RL1, RL2).



4.

Amplifier in accordance with claim 3,

characterized in that

the regulating device (B1)
comprises a first regulating entity (B2) to which as actual value a signal (avg), dependent
on the direct current voltage of the wo
rking points of first and second output (RL1, RL2),
and as desired value a predetermined direct current voltage level (V5) are applied.



5.

Amplifier in accordance with claim 3 or 4,

characterized in that

the regulating device
(B1) comprises a second regu
lating entity (B3) to which as actual value the voltage
-
difference (diff) between the first and second output (RL1, RL2) is applied whereat the
zero value of the voltage
-
difference acts as predetermined value.



6.

Amplifier in accordance with claim 5,

characterized in that

the regulating device (B1)
comprises a third regulating entity (B4), so that the predetermined direct current voltage
level is composed of the average potential value of the first and second input (+in,
-
in) by
means of a filter
device.



7.

Amplifier in accordance with claim 5,

characterized in that

the predetermined direct
current voltage level of the first regulating entity (B2) is composed, by means of a filter
device (R26, C3), of the average potential value (avgin) of those
inputs of the first and
second preamplifier blocks (N1B, N2B) which are each connected via a feedback network
with the first and second output (RL1, RL2) respectively.



8.

Amplifier in accordance with at least one of the claims 1 to 7,
characterized in tha
t

the
quiescent current control device (Ix) sets the quiescent current of the first and second final
amplifier (M1, M2) by means of a temperature dependent control value in which the
temperature is especially provided in the area of the final amplifier.



9.

Amplifier in accordance with at least one of the claims 1 to 7,
characterized in that

the
quiescent current control device (Ix) comprises a regulator which measures the actual
value of the quiescent current of at least one final amplifier (M1, M2) and re
adjusts the
quiescent current to a desired value.



10.

Amplifier in accordance with at least one of the claims 1 to 8,
characterized in
that

local feedback and/or compensation methods in both signal chains are applied to
enhance the linear transmission beh
avior.